Comparator

ABSTRACT

A comparator includes: a wide-swing operation transconductance amplifier (OTA), having first and second differential input pairs for receiving first and second differential input signals respectively, the wide-swing OTA generating first and second intermediate output voltages in comparing the first with the second differential input signals; a current switch group; a current mirror group, wherein when an input common mode voltage of the first and the second differential input signal tends to one of a first and a second reference voltage, one of the first and the second differential input pair is turned off, and the current switch group and the current mirror group compensate a current flowing through the other of the first and the second differential input pair; and a decision circuit coupled to the wide-swing OTA, for enlarging a voltage difference between the first and the second intermediate output voltage to output a voltage comparison output signal.

This application claims the benefit of Taiwan application Serial No.100104143, filed Feb. 8, 2011, the subject matter of which isincorporated herein by reference.

TECHNICAL FIELD

The disclosure relates in general to a comparator with dual differentialinput pairs, and more particularly to a comparator with a rail-to-railinput voltage range.

BACKGROUND

A comparator (also referred as ‘a voltage comparator’) is an integratedcircuit. The voltage comparator compares the magnitudes of two inputvoltages to determine which is higher. Which input voltage is higher isdetermined and indicated by an output voltage from the comparator.

As for the conventional comparators, when the conventional comparatorcompares the differential input voltages, the range of the input commonmode voltage identified by the comparator is not rail-to-rail. That is,the range of the input common mode voltage identified by the comparatoris not from the ground voltage GND to the operation voltage VDD. If theinput common mode voltage tends to the ground voltage GND, thecomparator with PMOS transistor differential input pair is used. To thecontrary, if the input common mode voltage tends to the operationvoltage VDD, the comparator with NMOS transistor differential input pairis used.

BRIEF SUMMARY OF THE DISCLOSURE

The disclosure is directed to a comparator with dual differential inputpairs. When the input common mode voltage tends to a ground voltage GND,the NMOS transistor differential input pair is turned off and a currentflowing through the PMOS transistor differential input pair iscompensated. When the input common mode voltage tends to an operationvoltage VDD, the PMOS transistor differential input pair is turned offand a current flowing through the NMOS transistor differential inputpair is compensated. Thus, the operation range of the input common modevoltage may be from the ground voltage GND to the operation voltage VDD,that is, rail-to-rail input.

According to an exemplary embodiment of the disclosure, a comparatorincluding a wide-swing operation transconductance amplifier (OTA), acurrent switch group, a current mirror group, and a decision circuit isprovided. The wide-swing OTA at least includes a first and a seconddifferential input pair for receiving a first and a second differentialinput signal respectively, and generates a first and a secondintermediate output voltage in comparing the first and the seconddifferential input signal. When an input common mode voltage of thefirst and the second differential input signal tends to one of a firstand a second reference voltage, one of the first and the seconddifferential input pair is turned off, and the current switch group andthe current mirror group compensate a current flowing through the otherof the first and the second differential input pair. The decisioncircuit enlarges a voltage difference between the first and the secondintermediate output voltage to output a voltage comparison outputsignal.

It is to be understood that both the foregoing general description andthe following detailed description are exemplary and explanatory onlyand are not restrictive of the disclosed embodiments, as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A, FIG. 1B and FIG. 2 are circuit diagrams of a rail-to-railcomparator according to an embodiment of the disclosure;

FIG. 3 shows a signal timing diagram of the embodiment of thedisclosure;

FIG. 4 shows a range of an input common mode voltage VCM of an OTA ofthe embodiment of the disclosure; and

FIG. 5A to FIG. 12B are circuit operation diagrams of the comparatoraccording to the embodiment of the disclosure.

DETAILED DESCRIPTION OF THE DISCLOSURE

FIG. 1A, FIG. 1B and FIG. 2 are circuit diagrams of a rail-to-railcomparator according to an embodiment of the disclosure. As indicated inFIG. 1A and FIG. 1B, the rail-to-rail comparator includes a wide-swingoperation transconductance amplifier (OTA), an input switch groupsw1˜sw4, NMOS capacitors M7C and M8C, current switches M21 and M22, afirst current mirror M19 and M20, a second current mirror M23 and M24,and a decision circuit 200. The rail-to-rail comparator receives a firstcontrol pulse signal ph1 and a second control pulse signal ph2.

The wide-swing OTA includes transistors M1˜M14 and switches sw5˜sw6. Thetransistors M1 and M2 form an NMOS transistor differential input pair.The transistors M3 and M5 form a current mirror. The transistors M4 andM6 form a current mirror.

The transistors M9 and M10 form a PMOS transistor differential inputpair. The transistors M11 and M13 form a current mirror. The transistorsM12 and M14 form a current mirror.

The transistors M19 and M20 form a current mirror. The transistor M21 isconnected to the NMOS transistor differential input pair M1 and M2 inparallel; and the transistor M21 and the current mirror M19 and M20together compensate a current flowing through the NMOS transistordifferential input pair M1 and M2.

The transistor M22 is connected to the PMOS transistor differentialinput pair M9 and M10 in parallel; and the transistor M22 and thecurrent mirror M23 and M24 together compensate a current flowing throughthe PMOS transistor differential input pair M9 and M10. The transistorsM23 and M24 form a current mirror.

Conduction of the switches sw1 and sw2 determine whether voltages VIPand VIN are coupled to nodes IPA and INA respectively. Conduction of theswitches sw3 and sw4 determine whether a voltage VDD/2 is coupled to thenodes IPA and INA respectively. The switch sw5 determines whether thegate and the drain of the transistor M7 are coupled to each other. Theswitch sw6 determines whether the gate and the drain of the transistorM8 are coupled to each other.

Referring to FIG. 2. The decision circuit 200 includes: transistorsM15˜M18, switches sw7˜sw12, Schmitt triggers 210 and 220, a logic gatecircuit 230 (such as but not limited to an XOR logic gate), a delaycircuit 240, a logic circuit 250 (such as but not limited to an ANDlogic gate) and a D-type flip-flop 260.

The transistors M15˜M18 form a positive feedback loop. The transistorsM15 and M18 are for raising the voltage VIP1 up to VDD or for loweringthe voltage VIN1 to the ground voltage GND. The transistors M16 and M17are for raising the voltage VIN1 up to VDD or for lowering the voltageVIP1 to the ground voltage GND.

Conduction of the switches sw7 and sw8 determine whether output voltagesVOUTP and VOUTN are coupled to the nodes IP1 and IN1 respectively.Conduction of the switches sw9 and sw10 determine whether the nodes IP1and IN1 respectively are coupled to the ground voltage GND. Conductionof the switches sw11 and sw12 respectively determines whether theoperation voltage VDD and the ground voltage GND are connected to thetransistors M15˜M18.

The Schmitt triggers 210 and 220 receive signals IP1 and IN1respectively, and output signals OUTP1 and OUTN1 respectively. The XORlogic gate 230 receives signals OUTP1 and OUTN1 and outputs a signals G.The delay circuit 240 delays the signal G by a delay time Td into asignal H. The AND logic gate 250 receives signals G and H and outputs asignal CK. The D-type flip-flop 260 determines whether to latch thesignal OUTP1 according to the signal CK.

FIG. 3 shows a signal timing diagram of the comparator of the embodimentof the disclosure. The control pulse signals ph1 and ph2 are inverse toeach other, and there is no overlapping between the high logic levels ofthe two signals. For better understanding, the waveforms of the signalsINA, OUTN, OUTN1 are marked by triangles (Δ) to be discriminated fromthe waveforms of the signals IPA, OUTP, OUTP1 (no marks).

FIG. 4 shows a range of an input common mode voltage VCM of the OTA. Thevoltage VCM is defined as: VCM=(VIPA+VINA)/2=(VIP+VIN)/2.

Within a first region, the input common mode voltage VCM of the OTA issmaller than VBN (VCM<VBN), wherein VBN is a bias voltage applied to theNMOS transistor M21. When the input common mode voltage VCM of the OTAis smaller than VBN, the average voltage of the gate voltages VIPA andVINA of the NMOS transistor differential input pair M1 and M2 is low sothat the node voltage VCMN is also low, making the gate-source voltageof the current switch M21 (i.e. VBN−VCMN) exceed its conductionthreshold voltage so that the current switch M21 is turned on, and thecurrent mirror M19 and M20 is also turned on. In addition, the averagevoltage of the gate voltage VIPA and VINA of the PMOS transistordifferential input pair M9 and M10 is low so that the node voltage VCMPis also low, making the source-gate voltage of the current switch M22(i.e. VCMP−VBP) is lower than its conduction threshold voltage, thecurrent switch M22 is turned off, and the current mirror M23 and M24 isalso turned off.

Within a second region, the input common mode voltage VCM of the OTAranges between VBN and VBP (VBN<VCM<VBP), wherein VBP is a bias voltageapplied to the PMOS transistors M22. When the input common mode voltageVCM of the OTA ranges between VBN and VBP, the average voltage of thegate voltages VIPA and VINA of the NMOS transistor differential inputpair M1 and M2 is moderate. Since the node voltage VCMN is not lowenough to make the gate-source voltage of the current switch M21 (thatis, VBN−VCMN) exceed its conduction threshold voltage, the currentswitch M21 is turned off. Since the node voltage VCMP is not high enoughto make the source-gate voltage of the current switch M22 (that is,VCMP−VBP) exceed its conduction threshold voltage, the current switchM22 is turned off. Within the second region, the current mirror M19 andM20 as well as the current mirror M23 and M24 are turned off.

Within a third region, the input common mode voltage VCM of the OTA islarger than bias voltage VBP. As the average voltage of the gatevoltages VIPA and VINA of the PMOS transistor differential input pair M9and M10 is high, the node voltage VCMP is also high, making thesource-gate voltage of the current switch M22 (that is, VCMP−VBP) exceedthe conduction threshold voltage of the current switch M22 so that thecurrent switch M22 is turned on, and the current mirror M23 and M24 isalso turned on. As the average voltage of the gate voltages VIPA andVINA of the NMOS transistor differential input pair M1 and M2 is high,the node voltage VCMN is also high, making the gate-source voltage ofthe current switch M21 (that is, VBN−VCMN) does not exceed theconduction threshold voltage of the current switch M21, so the currentswitch M21 is turned off and the current mirror M19 and M20 is alsoturned off.

The operations of the comparator according to the embodiment of thedisclosure are disclosed below, which include two steps. In the firststep, the first control pulse signal ph1 is at high logic level, thesecond control pulse signal ph2 is at low logic level, and thecomparator is equalized. In the second step, the first control pulsesignal ph1 is at low logic level and the second control pulse signal ph2is at high logic level so that the comparator performs a voltagecomparison and outputs a comparison result.

First Step: Equalization

Referring to FIG. 5A, FIG. 5B and FIG. 6, circuit operation diagrams ofthe comparator in the first step are shown. As indicated in FIG. 5A andFIG. 5B, since the first control pulse signal ph1 is at high logic leveland the second control pulse signal ph2 is at low logic level, theswitches sw3 and sw4 of the input switch group are turned on; the switchsw1 and sw2 are disconnected, and the switches sw5 and sw6 of the OTAare turned on. So, the comparator input voltages IPA and INA are bothequal to VDD/2, and the input common mode voltage VCM of the OTA rangesbetween VBN and VBP (VBN<VCM<VBP). That is, the input common modevoltage VCM is within the second region of FIG. 4. Thus, the currentswitches M21 and M22, the current mirror M19 and M20, and the currentmirror M23 and M24 are all turned off. In the first step, the gate biasvoltages VN7C and VN8C of the transistors M7 and M8 are established andstored in the NMOS capacitors M7C and M8C respectively. The steadycurrent of the OTA is indicated in FIG. 5A and FIG. 5B. In the firststep, the currents flowing through the NMOS transistor differentialinput pair M1 and M2, the PMOS transistor differential input pair M9 andM10, the current mirror M11 and M13, and the current mirror M12 and M14are all equal to I/2. The currents flowing through the current mirrorsM3 and M5, M4 and M6, and M7 and M8 are all equal to I.

Referring to FIG. 6. In the first step, in the decision circuit 200, theswitches sw9 and sw10 are turned on, and the switches sw7, sw8, sw11 andsw12 are disconnected, so the node voltages VIP1 and VIN1 of the nodesIP1 and IN1 are all equal to GND (0V). Thus, the outputs OUTP1 and OUTN1of the Schmitt trigger 210 and 220 are both at low logic level, theoutput G of the XOR logic gate 230 is at low logic level, and the outputCK of the AND logic gate 250 is at low logic level. Since the inputpulse signal CK of the D flip-flop 260 is at low logic level, the Dflip-flop 260 outputs a previous output data Q(n−1).

Second Step:

In the second step, the input switches sw3 and sw4 are disconnected, theswitches sw1 and sw2 are turned on, and the switches sw5 and sw6 of theOTA are disconnected. Meanwhile, the inputs IPA and INA are respectivelyconnected to the differential inputs IP and IN. Based on the magnitudeof the voltage of the input common mode voltage VCM of the OTA, thesecond step is further divided into 5 scenarios, namely, VBN<VCM<VBP,VDD≧VCM>VBP, VCM=VBP, VCM=VBN and GND≦VCM<VBN. The operations of thecomparator under the 5 scenarios are disclosed below.

Scenario 1 of the Second Step: VBN<VCM<VBP

If the voltage VIP of the differential input IP of the comparator isVCM+(ΔV/2), the voltage VIN of the differential input IN is VCM−(ΔV/2),and VBN<VCM<VBP, then the input common mode voltage of the OTA fallswithin the second region of FIG. 4. As disclosed above, within thesecond region, the current switches M21 and M22, the current mirror M19and M20, and the current mirror M23 and M24 are all turned off.

FIG. 7A, FIG. 7B and FIG. 8 are circuit operation diagrams of thecomparator in the second step and VBN<VCM<VBP. Referring to FIG. 7A andFIG. 7B. In the present scenario, the currents flowing through the PMOStransistor differential input pair M9 and M10 are I/2−(ΔI/2) andI/2+(ΔI/2) respectively, and the current flowing into the NMOStransistor differential input pair M1 and M2 are I/2+(ΔI/2) andI/2−(ΔI/2) respectively. The current flowing to the transistor M11 isI/2−(ΔI/2), and the current flowing to the transistor M12 is I/2+(ΔI/2).The transistors M11 and M13 form a current mirror, so the currentsflowing through the two transistors are the same. Likewise, thetransistors M12 and M14 form a current mirror, so the currents flowingthrough the two transistors are the same. The total current flowingthrough the transistor M3 is the sum of the two currents flowing throughthe transistors M1 and M14; and the total current flowing through thetransistor M4 is the sum of the two currents flowing through thetransistors M2 and M13. Thus, the total currents flowing through thetransistors M3 and M4 are I+ΔI and I−ΔI respectively. Since thetransistors M3 and M5 form a current mirror, the current flowing throughthe transistors M5 is I+ΔI. Since the transistors M4 and M6 form acurrent mirror, the current flowing through the transistor M6 is I−ΔI.

Since the gate voltages of the transistor M7 and M8 are already storedin the NMOS capacitors M7C and M8C respectively in the first step, thecurrents flowing through the transistor M7 and M8 are I. The currentflowing to the node OUTP from the transistor M5 is I+ΔI, but the currentflowing to the transistor M7 from the node OUTP is I, and the currentdifference ΔI flows to the decision circuit 200 via the node OUTP. Dueto parasitic capacitance effect, the current difference ΔI causes thevoltage VOUTP of the node OUTP to be raised. Similarly, the currentflowing to the node OUTN from the transistors M6 is I−ΔI, but thecurrent flowing to the transistor M8 from the node OUTN is I, and thecurrent difference ΔI is provided by the decision circuit 200 (i.e. thedecision circuit 200 provides the current difference ΔI to thetransistor M8 via the node OUTN). Due to parasitic capacitance effect,the current difference ΔI causes the voltage VOUTN of the node OUTN tobe lowered.

Referring to FIG. 8. The switches sw9 and sw10 of the decision circuit200 are disconnected, the switches sw7 and sw8 are turned on so that thenodes IP1 and IN1 are connected to the node OUTP and OUTN respectively.The switch sw12 is turned on so that the sources of the transistors M15and M16 are connected to the operation voltage VDD via the switch sw12.The switch sw11 is turned on so that the sources of the transistors M17and M18 are connected to the ground voltage GND via the switch sw11. Asdisclosed above, the voltage VOUTP of the node OUTP is raised so thatthe voltage VIP1 of the nodes IP1 is raised as well; and the voltageVOUTN of the node OUTN is lowered so that the voltage VIN1 of the nodeIN1 is lowered as well. As the voltage VIP1 is raised, the gate-sourcevoltage of the transistors M18 is higher, increasing the current flowingthrough the transistors M18 and lowering the voltage VIN1 of the nodesIN1. Similarly, as the voltage VIN1 of the node IN1 is lowered, thegate-source voltage of the transistor M17 is reduced, decreasing thecurrent of the transistor M17 and raising the voltage VIP1 of the nodeIP1. Thus, the transistors M15˜18 form a positive feedback loop(designated by the dotted lines of FIG. 8), making the voltages VIP1 andVIN1 towards different directions. That is, the voltage VIP1 graduallytends to the operation voltage VDD, and the voltage VIN1 gradually tendsto the ground voltage GND. The outputs OUTP1 and OUTN1 of the Schmitttrigger 210 and 220 are high logic level and low logic levelrespectively, so the output G of the XOR logic gate 230 transits to highlogic level from low logic level. The delay time of the delay circuit240 is Td. After the time Td (from the transition of the output G of theXOR logic gate 230), the output CK of the AND logic gate 250 transits tohigh logic level from low logic level. After the signal CK transits tohigh logic level, the D-type flip-flop 260 latches the H signal OUTP1,so the output Q(n) of the D-type flip-flop 260 is at high logic level 1.Q(n)=1 indicates that the comparator detects and determines that theinput voltage VIP is larger than the input voltage VIN.

Similarly, if the voltage VIP of the differential input IP of thecomparator is VCM−(ΔV/2), the voltage VIN of the differential input INis VCM+(ΔV/2), and VBN<VCM<VBP, then the output Q(n) of the D-typeflip-flop 260 is at low logic level 0. Q(n)=0 indicates that thecomparator detects and determines that the input voltage VIP is smallerthan the input voltage VIN (due to the operations of the decisioncircuit 200, the voltage VIP1 gradually tends to the ground voltage GND,and the voltage VIN1 gradually tends to the operation voltage VDD).

Scenario 2 of the Second Step: VDD≧VCM>VBP

FIG. 9A and FIG. 9B are circuit operation diagrams of the comparator inthe second step and VDD≧VCM>VBP. If the voltage VIP of the differentialinput IP of the comparator is VCM+(ΔV/2), the voltage VIN of thedifferential input IN is VCM−(ΔV/2), and VCM>VBP, then the input commonmode voltage of the OTA falls within the third region. As disclosedabove, the current switch M21 and the current mirror M19 and M20 areturned off, but the current switch M22 and the current mirror M23 andM24 are turned on, as indicated in FIG. 9A and FIG. 9B.

In this scenario, as the average voltage of the gate voltages VIPA andVINA of the PMOS transistor differential input pair M9 and M10 is high,the node voltage VCMP is raised accordingly, and the transistors M9˜14are turned off. Since the current I flows into the drain of thetransistor M23, the current flowing through the transistor M23 isprovided by the NMOS transistor differential input pair M1 and M2. Thecurrent flowing through the node CMN changes to 2I from I, wherein acurrent I flows to the current source I from the node CMN, and a currentI flows to the transistor M23 from the node CMM. Given that VDD≧VCM>VBP,the current flowing through the node CMN is doubled (in comparison tothe scenario that VBN<VCM<VBP), so the currents flowing to the NMOStransistor differential input pair M1 and M2 are doubled as I+ΔI andI−ΔI from I/2+(ΔI/2) and I/2−(ΔI/2) respectively, and the total currentsflowing through the transistors M3 and M4 are I+ΔI and I−ΔIrespectively. Due to current mirroring, the currents flowing through thetransistors M5 and M6 are I+ΔI and I−ΔI respectively. Since the gatevoltages of the transistors M7 and M8 are stored in the NMOS capacitorsM7C and M8C respectively in the first step, the currents flowing throughthe transistor M7 and M8 still remain at I.

The current flowing to the node OUTP from the transistors M5 is I+ΔI,but the current flowing to the transistor M7 from the node OUTP is I.The current difference ΔI flows to the decision circuit 200 via the nodeOUTP. Due to parasitic capacitance effect, the current difference ΔIcauses the voltage VOUTP of the node OUTP to be raised. Similarly, thecurrent flowing to the node OUTN from the transistor M6 is I−ΔI, but thecurrent flowing to the transistor M8 from the node OUTN is I. A currentdifference ΔI is provided by the decision circuit 200. Due to parasiticcapacitance effect, the current difference ΔI causes the voltage VOUTNof the node OUTN to be lowered. In this scenario, the operations of thedecision circuit 200 are identical or similar to FIG. 8, and the detailsare not repeated here. Q(n)=1 indicates that the comparator detects anddetermines that the input voltage VIP is larger than the input voltageVIN. Q(n)=0 indicates that the comparator detects and determines thatthe input voltage VIP is smaller than the input voltage VIN.

Scenario 3 of the Second Step: VCM=VBP

FIG. 10A and FIG. 10B are circuit operation diagrams of the comparatorin the second step and VCM=VBP. If the voltage VIP of the differentialinput IP of the comparator is VCM+(ΔV/2), the voltage VIN of thedifferential input IN is VCM−(ΔV/2) and VCM=VBP, then the input commonmode voltage of the OTA falls on the border between the second regionand the third region. The current switch M21 and current mirror M19 andM20 are turned off, but the current switch M22 and the current mirrorM23 and M24 are turned on. If the aspect ratio (the ratio of the channelwidth to the channel length) of the transistor M22 is the same as thesum of the aspect ratios of the transistors M9 and M10, then the currentflowing to the transistor M22 is I/2, and the current flowing to thedrain of the transistor M23 is also I/2.

Since the transistor M23 drains a current I/2 from the node CMN, thecurrent flowing through the node CMN become 1.5 times (in comparison tothe scenario when VBN<VCM<VBP). Thus, the currents flowing to the NMOStransistor differential pair M1 and M2 become 1.5 times (in comparisonto the scenario when VBN<VCM<VBP), as I*¾+(ΔI*¾) and I*¾−(ΔI*¾) fromI/2+(ΔI/2) and I/2−(ΔI/2) respectively.

Likewise, since the transistor M22 drains a current I/2 from the nodeCMP, the transistor M22 drains away a half of the current provided tothe node CMP by the current source I (that is, I/2 is drained by thetransistor M22). Consequently, the currents flowing to the PMOStransistor differential input pair M9 and M10 are halved (in comparisonto the scenario when VBN<VCM<VBP), and the currents flowing through theM10 and M9 are halved as to I/4+(ΔI/4) and I/4−(ΔI/4) from I/2+(ΔI/2)and I/2−(ΔI/2) respectively.

The current flowing into the transistor M11 is the same as the currentflowing through the transistor M9, i.e. I/4−(ΔI/4). The current flowingto the transistor M12 is the same as the current flowing through thetransistor M10, i.e. I/4+(ΔI/4). Due to current mirroring, the currentsflowing to the transistors M13 and M14 are I/4−(ΔI/4) and I/4+(ΔI/4)respectively. Thus, the total currents flowing through the transistorsM3 and M4 are I+ΔI and I−ΔI respectively. Due to current mirroring, thetotal currents flowing through the transistors M5 and M6 are I+ΔI andI−ΔI respectively. Since the gate voltages of the transistors M7 and M8are stored in the NMOS capacitors M7C and M8C respectively in the firststep, the currents flowing through the transistor M7 and M8 still remainat I.

The current flowing to the node OUTP from the transistor M5 is I+ΔI, butthe current flowing to the transistor M7 from the node OUTP is I. So, acurrent difference ΔI flows to the decision circuit 200 via the nodeOUTP. Due to parasitic capacitance effect, the current difference ΔIcauses the voltage VOUTP of the node OUTP to be raised. Similarly, thecurrent flowing to the node OUTN from the transistor M6 is I−ΔI, but thecurrent flowing to the transistor M8 from the node OUTN is I. A currentdifference ΔI is provided by the decision circuit 200. Due to parasiticcapacitance effect, the current difference ΔI causes the voltage VOUTNof the node OUTN to be lowered. In this scenario, the operations of thedecision circuit 200 are identical or similar to FIG. 8, and the detailsare not repeated here. Q(n)=1 indicates that the comparator detects anddetermines that the input voltage VIP is larger than the input voltageVIN. Q(n)=0 indicates that the comparator detects and determines thatthe input voltage VIP is smaller than the input voltage VIN.

Scenario 4 of the Second Step: VCM=VBN

FIG. 11A and FIG. 11B are circuit operation diagrams of the comparatorin the second step and VCM=VBN. If the voltage VIP of the differentialinput IP of the comparator is VCM+(ΔV/2), the voltage VIN of thedifferential input IN VCM−(ΔV/2), and VCM=VBN, then the input commonmode voltage of the OTA falls on the border between the first region andthe second region. The current switch M22 and the current mirror M23 andM24 are turned off, but the current switch M21 and the current mirrorM19 and M20 are turned on. If the aspect ratio of the transistor M22 isthe same as the sum of the aspect ratios of the transistors M1 and M2,then the current flowing through the transistor M21 is I/2, and thecurrent flowing through the drain of the transistor M20 is also I/2.

Since the transistor M21 provides a current I/2 to the node CMN, thecurrent provided to the node CMN by the NMOS transistor differentialinput pair M1 and M2 is halved (that is, I/2) in comparison to thescenario VBN<VCM<VBP. Thus, the currents flowing to the NMOS transistordifferential pair M1 and M2 is halved (in comparison to the scenarioVBN<VCM<VBP) as I/4+(ΔI/4) and I/4−(ΔI/4) from I/2+(ΔI/2) and I/2−(ΔI/2)respectively.

Likewise, since the transistor M20 provides a current I/2 to the nodeCMP and the current source still provides a current I to the node CMP,the currents flowing to the PMOS transistor differential input pair M9and M10 become 1.5 times (in comparison to the scenario VBN<VCM<VBP).Thus, the currents flowing through the M10 and M9 become 1.5 times, asI*¾+(ΔI*¾) and I*¾−(ΔI*¾) from I/2+(ΔI/2) and I/2−(ΔI/2) respectively.

The current flowing to the transistor M11 is the same as the currentflowing through the transistor M9, i.e. I*¾−(ΔI*¾). The current flowingto the transistor M12 is the same as the current flowing through thetransistor M10, i.e. I*¾+(ΔI*¾). Due to current mirroring, the currentsflowing to the transistors M13 and M14 are I*¾−(ΔI*¾) and I*¾+(ΔI*¾)respectively. Thus, the total currents flowing through the transistorsM3 and M4 are I+ΔI and I−ΔI respectively. Due to current mirroring, thetotal currents flowing through the transistors M5 and M6 are I+ΔI andI−ΔI respectively. Since the gate voltages of the transistors M7 and M8are stored in the NMOS capacitors M7C and M8C in the first steprespectively, the currents flowing through the transistor M7 and M8still remain at I.

The current flowing to the node OUTP from the transistor M5 is I+ΔI, butthe current flowing to the transistor M7 from the node OUTP is I. Thecurrent difference ΔI flows to the decision circuit 200 via the nodeOUTP. Due to parasitic capacitance effect, the current difference ΔIcauses the voltage VOUTP of the node OUTP to be raised. Similarly, thecurrent flowing to the node OUTN from the transistor M6 is I−ΔI, but thecurrent flowing to the transistor M8 from the node OUTN is I. Thecurrent difference ΔI is provided by the decision circuit 200. Due toparasitic capacitance effect, the current difference ΔI causes thevoltage VOUTN of the node OUTN to be lowered. In this scenario, theoperations of the decision circuit 200 are identical or similar to FIG.8, and the details are not repeated here. Q(n)=1 indicates that thecomparator detects and determines that the input voltage VIP is largerthan the input voltage VIN. Q(n)=0 indicates that the comparator detectsand determines that the input voltage VIP is smaller than the inputvoltage VIN.

Scenario 5 of the Second Step: GND≦VCM<VBN

FIG. 12A and FIG. 12B are circuit operation diagrams of the comparatorin the second step and GND≦VCM<VBN. If the voltage VIP of thedifferential input IP of the comparator is VCM+(ΔV/2), the voltage VINof the differential input IN is VCM−(ΔV/2), and GND≦VCM<VBN, then theinput common mode voltage of the OTA falls within the first region. Thecurrent switch M21 and the current mirror M19 and M20 are turned on, butthe current switch M22 and the current mirror M23 and M24 are turnedoff. The NMOS transistor differential input pair M1 and M2 is turnedoff.

Since the transistor M20 and the current source both provide currents Ito the node CMP, the current flowing through the node CMP is doubled (incomparison to the scenario when VBN<VCM<VBP). Thus, the currents flowingthrough the PMOS transistor differential input pair M10 and M9 isdoubled as I+ΔI and I−ΔI from I/2+(ΔI/2) and I/2−(ΔI/2) respectively.

The current flowing to the transistor M11 is the same as the currentflowing through the transistor M9, i.e. I−ΔI. The current flowing to thetransistor M12 is the same as the current flowing through the transistorM10, i.e. I+ΔI. Due to current mirroring, the currents flowing to thetransistors M13 and M14 are I−ΔI and I+ΔI respectively. Thus, the totalcurrents flowing through the transistors M3 and M4 are I+ΔI and I−ΔIrespectively. Due to current mirroring, the total currents flowingthrough the transistors M5 and M6 are I+ΔI and I−ΔI respectively. Sincethe gate voltages of the transistors M7 and M8 are stored in the NMOScapacitors M7C and M8C in the first step respectively, the currentsflowing through the transistor M7 and M8 still remain at I.

The current flowing to the node OUTP from the transistor M5 is I+ΔI, butthe current flowing to the transistor M7 from the node OUTP is I. Thecurrent difference ΔI flows to the decision circuit 200 via the nodeOUTP. Due to parasitic capacitance effect, the current difference ΔIcauses the voltage VOUTP of the node OUTP to be raised. Similarly, thecurrent flowing to the node OUTN from the transistor M6 is I−ΔI, but thecurrent flowing to the transistor M8 from the node OUTN is I. Thecurrent difference ΔI is provided by the decision circuit 200. Due toparasitic capacitance effect, the current difference ΔI causes thevoltage VOUTN of the node OUTN to be lowered. In this scenario, theoperations of the decision circuit 200 are identical or similar to FIG.8, and the details are not repeated here. Q(n)=1 indicates that thecomparator detects and determines that the input voltage VIP is largerthan the input voltage VIN. Q(n)=0 indicates that the comparator detectsand determines that the input voltage VIP is smaller than the inputvoltage VIN.

To summarize, in the above embodiment of the present disclosure, if theinput common mode voltage VCM tends to the ground voltage GND (forexample, within the first region of FIG. 4), then the NMOS transistordifferential input pair is turned off, but the current flowing to thePMOS transistor differential input pair is compensated. As indicated inFIG. 12A and FIG. 12B, the compensation current is provided by thecurrent mirror M19 and M20. Likewise, if the input common mode voltageVCM tends to the operation voltage VDD (for example, within the thirdregion of FIG. 4), the PMOS transistor differential input pair is turnedoff, but the current flowing to the NMOS transistor differential inputpair is compensated. As indicated in FIG. 9A and FIG. 9B, thecompensation current is provided by the current mirror M23 and M24 (infact, the transistor M23 drains a current, and the NMOS transistordifferential input pair provides higher currents). By compensatingcurrents flowing to the differential input pair, the voltages VOUTP andVOUTN are quickly changed toward opposite directions (one is raised tothe operation voltage VDD, and the other is lowered to the groundvoltage GND) and the comparison results is obtained promptly. So thatthe range of the input common mode voltage identified by the comparatorcircuit according to the embodiment of the disclosure is rail-to-rail.

It will be appreciated by those skilled in the art that changes could bemade to the disclosed embodiments described above without departing fromthe broad inventive concept thereof. It is understood, therefore, thatthe disclosed embodiments are not limited to the particular examplesdisclosed, but is intended to cover modifications within the spirit andscope of the disclosed embodiments as defined by the claims that follow.

What is claimed is:
 1. A comparator, comprising: a wide-swing operationtransconductance amplifier (OTA), at least comprising a first and asecond differential input pair for receiving a first and a seconddifferential input signal respectively, the wide-swing OTA generating afirst and a second intermediate output voltage in comparing the firstand the second differential input signal; a current switch group; acurrent mirror group, wherein when an input common mode voltage of thefirst and the second differential input signal tends to one of a firstand a second reference voltage, one of the first and the seconddifferential input pair is turned off, and the current switch group andthe current mirror group compensate a current flowing through the otherof the first and the second differential input pair; and a decisioncircuit coupled to the wide-swing OTA, for enlarging a voltagedifference between the first and the second intermediate output voltageto output a voltage comparison output signal.
 2. The comparatoraccording to claim 1, further comprising: an input switch group coupledbetween the wide-swing OTA and the first and the second differentialinput signal, for selectively conducting the first and the seconddifferential input signal to the wide-swing OTA.
 3. The comparatoraccording to claim 1, further comprising: a plurality of voltagemaintain circuits coupled to the wide-swing OTA, for maintaining aplurality of internal gate bias voltages of the wide-swing OTA.
 4. Thecomparator according to claim 1, wherein, when the comparator isoperated in an equalization stage under control of a plurality ofcontrol pulse signals, the first and the second differential input pairare turned on, two currents flowing through the first and the seconddifferential input pair are the same, the first and the secondintermediate output voltage have the same level, and the decisioncircuit outputs a previous voltage comparison output signal.
 5. Thecomparator according to claim 1, wherein, the current switch groupcomprises a first and a second current switch; and the current mirrorgroup comprises a first and a second current mirror coupled to the firstand the second current switch respectively.
 6. The comparator accordingto claim 5, wherein, the first and the second reference voltage comprisea ground voltage and an operation voltage; when the common mode voltageranges between a first bias voltage and a second bias voltage, the firstcurrent switch and the first current mirror are turned off, the secondcurrent switch and the second current mirror are turned off, the firstand the second differential input pair are turned on, and a currentdifference flows between the wide-swing OTA and the decision circuit sothat one of the first and the second intermediate output voltage israised and the other is lowered, and the decision circuit forms apositive feedback loop to further enlarge the voltage difference betweenthe first and the second intermediate output voltage so that one of thefirst and the second intermediate output voltage is raised to theoperation voltage and the other is lowered to the ground voltage.
 7. Thecomparator according to claim 6, wherein, when the common mode voltageis higher than the second bias voltage and tends to the operationvoltage, the first current switch and the first current mirror areturned off, the second current switch and the second current mirror areturned on, the first and the second differential input pair are turnedon and off respectively, the second current switch and the secondcurrent mirror compensate a first differential current flowing throughthe first differential input pair, the current difference flowingbetween the wide-swing OTA and the decision circuit so that one of thefirst and the second intermediate output voltage is raised and the otheris lowered, and the decision circuit forms a positive feedback loop tofurther enlarge the voltage difference between the first and the secondintermediate output voltage so that one of the first and the secondintermediate output voltage is raised to the operation voltage and theother is lowered to the ground voltage.
 8. The comparator according toclaim 7, wherein, when the common mode voltage is the same as the secondbias voltage, the first current switch and the first current mirror areturned off, the second current switch and the second current mirror areturned on, the first and the second differential input pair are turnedon, the second current switch and the second current mirror compensatethe first differential current flowing through the first differentialinput pair, the current difference flows between the wide-swing OTA andthe decision circuit so that one of the first and the secondintermediate output voltage is raised and the other is lowered, and thedecision circuit forms a positive feedback loop to further enlarge thevoltage difference the first and the second intermediate output voltageso that one of the first and the second intermediate output voltage israised to the operation voltage and the other is lowered to the groundvoltage.
 9. The comparator according to claim 8, wherein, when thecommon mode voltage is the same as the first bias voltage, the firstcurrent switch and the first current mirror are turned on, the secondcurrent switch and the second current mirror are turned off, the firstand the second differential input pair are turned on, the first currentswitch and the first current mirror compensate a second differentialcurrent flowing through the second differential input pair, the currentdifference flows between the wide-swing OTA and the decision circuit sothat one of the first and the second intermediate output voltage israised and the other is lowered, and the decision circuit forms apositive feedback loop to further enlarge the voltage difference betweenthe first and the second intermediate output voltage so that one of thefirst and the second intermediate output voltage is raised to theoperation voltage and the other is lowered to the ground voltage. 10.The comparator according to claim 9, wherein, when the common modevoltage is lower than the first bias voltage and tends to the groundvoltage, the first current switch and the first current mirror areturned on, the second current switch and the second current mirror areturned off, the first and the second differential input pair are turnedoff and on respectively, the first current switch and the first currentmirror compensate the second differential current flowing through thesecond differential input pair, the current difference flows between thewide-swing OTA and the decision circuit so that one of the first and thesecond intermediate output voltage is raised and the other is lowered,and the decision circuit forms a positive feedback loop to furtherenlarge the voltage difference the first and the second intermediateoutput voltage so that one of the first and the second intermediateoutput voltage is raised to the operation voltage and the other islowered to the ground voltage.